This invention relates to an electric energy power converter comprising a unidirectional voltage source and a voltage inverter circuit arranged to convert said unidirectional voltage into an alternative voltage, possibly polyphase, intended to supply an inductive load, such as an electric motor.
In their industrial utilization, the electric energy power converters are mainly used for the supply of inductive loads such as electric motors. Presently it is of maximum interest to supply the electric motors by alternating current. Taking into consideration the voltage supplied circuits, the generation of the alternating voltages intended to supply the electric motors is obtained by modulating and inverting a direct supply voltage (or a unidirectional voltage, namely a voltage having a non-null medium value and always the same direction), which on its turn is usually obtained by converting an alternating voltage.
FIG. 1 of the accompany drawings shows the principle circuit diagram for generating variable three-phase alternating voltages applied to an inductive load such as an electric motor ME, starting from a source E t of a unidirectional volta v.sub.bus. The inverter INV is formed by a number of so-called "legs", equal to the number of phases to be generated, and therefore there are three legs G1, G2, G3 for a three-phase inverter. Each leg of the inverter is embodied by using valves formed by semiconductors which perform the ideal function of a bidirectional switch, although the components per se are unidirectional such as diodes and transistors, and it is controlled by a control circuit which imposes the transistor commutations (forced switching). The control circuit is CC.
FIG. 2 shows a typical embodiment of an inverter leg using two transistors T+, T- and two recirculation diodes D+, D-. By disregarding the voltage drops across the semiconductors, the voltage v.sub.u applied to the load may have only two instant values, namely O or v.sub.bus, irrespective of the direction (sign) of current i.sub.u which passes through the load, which we will consider as positive when it is directed according to the arrow shown in FIG. 2. When i.sub.u is positive, T+ should be conducting in order to have v.sub.u =v.sub.bus, otherwise v.sub.u =O due to the conduction through the recirculation diode D-. Therefore, by adequately operating transistor T+, the mean value of the output voltage v.sub.u may be controlled at will between O and V.sub.bus, by modulating the periods of duration of the two instant values O and v.sub.bus, with a commutation frequency suitably high in order to limit the current waves produced by the two-values wave form of voltage v.sub.u.
The technological problems posed by the embodiment of such an inverter circuit with forced switching are due to the high commutation frequency (typically more than 1 kHz), because to each commutation of the output voltage correspond a higher energy to be dissipated and a heavier electrical stress of the semiconductors. The energy lost by the commutation during a complete period, multiplied by the commutation frequency, gives a mean value of lost power, which imposes a limit to the maximum allowable commutation frequency; often, the commutation frequency is chosen by a compromise, by sharing out the conduction dissipation and the commutation dissipation.
Moreover, the electric stress which arises during the commutation imposes a poor exploitation of the components, in order to ensure the circuit a suitable reliability. FIG. 3 shows a typical locus v-i (diagram voltage-current) for a transistor used in a circuit with forced switching. The area S wherein the operation is reliable is ideally determined by the maximum voltage V.sub.max and by the maximum current I.sub.max which the transistor may withstand; Ap indicates the opening condition of the transistor and Co the conduction state thereof; line 1 is the diagram of the operating conditions of the transistor during a transition from conduction to opening, and line 2 is the diagram of the operating conditions of the transistor during a transition from opening to conduction. It may be remarked that, in order that the instant operating conditions are always contained within the area S, wherein the operating conditions are reliable, the maximum voltage V.sub.A which may be applied to the transistor during the opening thereof is noticeably lower than V.sub.max, and the maximum current I.sub.c which may pass through the transistor during the conduction thereof is noticeably lower than I.sub.max.
In addition to the above, in the case of short circuit on the load it is likely that the current limits of the transistors are overcome, and advanced systems for diagnostics and real time intervention are needed in order to prevent such a case.
In any event, the simultaneous existence of the three above factors imposes a very poor exploitation of the semiconductors, or, alternatively, a dangerous vulnerability of the circuit with respect to the fault and/or the transitory overcharge conditions should be accepted.
Different circuit configurations are known, which aid the semiconductors during the commutations, but these solutions need reactive bipoles, with the consequent encumbrances and costs, and complicated modulation algorithms; however, they do not qualitatively solve the problem, namely they do not allow a better exploitation of the semiconductors.
Moreover, the forced switching circuits do not allow use of thyristors, which are more sturdy than the transistors with respect to the operation in the conduction and opening statuses, but are able to compute from the conduction to the opening only in the absence of a current flowing through them.
A kind of inverter circuit, well different from the forced switching circuit pointed out above, is offered by the "resonant bus" technique, according to which the supply volt v.sub.bus applied to the inverter is a high frequency voltage oscillation (which is unidirectional for the power applications), obtained by means of a L-C resonator having a prevailing power. FIG. 4 shows the principle diagram of a power converter with resonant bus, supplied by a direct voltage source E, with a series resonant circuit L-C and a non-null mean voltage. In this case the legs of the inverter INV are free from commutation problems, because at each oscillation period the voltage v.sub.bus passes by zero; by suitably choosing the time intervals for doing the commutations, there is on the semiconductors no excessive stress. In such a kind of circuit the use of the gate turnoff thyristors (GTO) is allowed, but even in such a circuit the usual thyristors may not be used, except by adding further circuits.
The sole problem posed by the valves of a resonant bus inverter attains to- the scarce exploitation with respect to the voltage, because in general the peak voltage value to which the valves are subjected is equal or more than twice the mean supply voltage, which is useful for the power generation. This is clarified by FIG. 5, which shows (in a manner similar to FIG. 3) a typical locus v-i for a power valve used in a resonant bus inverter. It is to be remarked that the peak voltage value V.sub.pk is twice the mean voltage V.sub.out, useful for the power generation. Therefore, even in this case, there is a noticeable under-exploitation of the semiconductors.
However the characteristic and heavier problems of a resonant bus inverter are posed by the dimensioning of the power resonator L-C and by the drive and control techniques relating thereto, which are put into effect by means of special power circuits, diagrammatically shown in FIG. 4 by the blocks SW1 and SW2. Some improvements to the condition of poor exploitation of the semiconductors may be attained only through further circuit complications in blocks SW1 and SW2.